1. Field of the Invention
Apparatuses and methods consistent with the present invention relate to a mobile communication system. More particularly, the present invention relates to a signal acquisition apparatus and method for reducing a false alarm rate when a receiver detects synchronization of received data.
2. Description of the Related Art
In a general mobile burst communication system, a receiving end should acquire code synchronization prior to the acquisition of received information. Typically, the acquisition of the code synchronization employs a matched filter or an adaptive correlator. At the end of the matched filter of the adaptive correlator, a detector is provided for determining the presence or absence of the synchronization.
The acquisition of the code synchronization is crucial to a spread spectrum receiver. The synchronization acquisition can be detected through an active scheme, a passive scheme, and a combination of the two schemes. The passive scheme utilizes a filter matched to the spreading code, that is, utilizes a matched filter. The output of the matched filter is input to a threshold detector which determines whether to synchronize. The selection of the threshold greatly affects the performance of the active and passive synchronization acquisitions.
A conventional spread spectrum communication system takes advantage of a PN code or a code modified somewhat from the PN code. For the synchronization acquisition of the PN code, an apparatus using an analog circuit or a digital circuit detects a time when a correlation value is maximum or exceeds a threshold by use of correlation characteristics of the PN code.
In case of a general burst transmission system, it is risky to determine a correlation peak as a maximum value of the output of the matched filter to acquire the synchronization of the received signal because a start point of the bursts are not known. Thus, after prescribing a proper threshold, the correlation peak is determined when the output of the matched filter exceeds the prescribed threshold.
The conventional synchronization acquisition method takes a correlation value between a PN code received during one cycle or a cycle in part of the PN code, and a PN code generated at the receiver. Accordingly, the correlation value is affected in proportion to an amount of errors occurring due to noise or interference over a channel. As a result, erroneous synchronization acquisition may be caused. In view of probability, a probability of the erroneous synchronization acquisition is not zero. An ideal correlation value differs from a substantial correlation value because of the occurrence of the error by the channel status. Hence, a determination may be made that no synchronization is acquired even when it is synchronized according to the threshold of the correlation value which determines the synchronization acquisition, which is called a missing probability. Conversely, the synchronization acquisition may be determined even when it is not substantially synchronized, which is called a false alarm probability. In this respect, the verification procedure of the synchronization circuit is required according to an applied system. That is, the error due to the channel conditions affects in proportion to the correlation value that estimates the synchronization acquisition, and therefore acts as a crucial requirement in the system design.
To respond this, several methods have been proposed to determine the correlation value for the effective synchronization acquisition. A method calculates intersections of the threshold during code periods, and increases or decreases the threshold so as to obtain a desired number of intersections. It is noted that this method requires several periods and is suitable only for a special case in the code acquisition.
Meanwhile, a variance of the noise is roughly calculated using linear finite impulse response (FIR) filters. The variance of the noise or the interference is predictable, but determination knowledge of the threshold is required with respect to the channel status. For example, a constant false alarm rate (CFAR) is determined using a Rayleigh scattering of the decision variable via an additive white Gaussian noise (AWGN) channel. At this time, a reference channel filter is required, which should be orthogonal to the intended channel. If the unbalance of the reference channel impulse response of the generated code does not match the impulse response of the spreading code, it is well adopted in the noise environment but may be problematic in a jamming environment such as CW jamming.
For example, such a problem occurs when a supplementary code pair orthogonal to the codes and a code pair are adopted. Typically, every method for acquiring a fixed false alarm probability demands prior knowledge of the channel status, such as noise variance, the number of other users in a code division multiple access (CDMA) system, and a jamming factor.
One of the conventional methods to determine the threshold, takes advantage of an incoherent matched filter code synchronization acquisition structure. The output signal of the matched filter is proportional to an autocorrelation function (ACF) of the spreading code. Sampling at the output of an envelope detector is conducted by at least chip rate. A comparator uses a threshold Th to determine whether the output is ‘0’ or ‘1’. When the threshold intersects the correlation value at zero delay, its output is detected. In contrast, when the threshold intersects at other delays, the false alarm is detected. A rank filter orders output samples of N-ary matched filters in an ascending order, and obtains the threshold from N (window length) and a predefined location K. The false alarm rate can be acquired based on Equation 1.
                              P          fa                =                              N            -            K            +            1                    N                                    [                  Equation          ⁢                                          ⁢          1                ]            
Provided that the threshold is calculated from K=N/2, its result is at the middle of the sample set and Pfa is about 0.5. The delay is a half of the sample set, that is, N/2 samples. A new threshold is given at the chip rate.
Specifically, a PN code for synchronization is input to the matched filter and compared bit by bit with a PN code for synchronization stored in the receiver. When the corresponding bits of the two codes are identical, ‘1’ is output, and when the corresponding bits are different, ‘0’ is output. Accordingly, when the two codes match in most bits, the greatest number of ‘1’s is output and the output values of the matched filter are converted to an energy value X through the envelope detector. In other words, when the PN codes are identical and completely match with time, a very high correlation value is produced and thus a maximum energy value is generated. Otherwise, a relative small energy value is generated since a very low correlation value is produced.
Besides the passive scheme using the matched filter in the synchronization circuit as discussed above, the active scheme employs a cross correlator. The use of the cross correlator can speed up the detection but is subject to the complex implementation in comparison with the matched filter. However, the two schemes produce the same result.
As mentioned above, the energy value can determine whether the PN synchronization matches using a set threshold.
However, at this time, it may be miscalculated that the synchronization is performed even when there is no desired code. A parameter relating to the miscalculation probability is the false alarm probability. A threshold determination method has been under development to fix the false alarm probability and maximize a detection probability.
One of the threshold determination methods is a fixed threshold determination method. This method fixes the threshold to an extracted value of the energy value of the background noise received at the initial code synchronization, and determines whether it is code synchronized. However, the fixed threshold method causes various problems not only in the code synchronization acquisition for every change of the background noise of the wireless channel environment but also over a multipath channel. To be brief, this method shows poor adaptability to the change of the channel environment.
Another method averages N-ary output values of a received code signal that has passed through a matched filter or a correlator, and utilizes the average value as the threshold. This method shows the optimal performance in a homogeneous wireless channel environment, that is, when the background noise is constant independently from the magnitude. However, when the background noise changes, the code synchronization time period may be lengthened or the false alarm rate may increase with respect to the received code. In addition, since arbitrary N-ary windows contain the multipath signal in the multipath channel, the threshold is increased and the code synchronization time period is lengthened.
Still another method processes the received code signal at the matched filter or the correlator of the receiving end, orders N-ary output values, and uses a value having a certain magnitude as a code synchronization threshold. This method shows excellent performance in the multipath channel but lengthens the code synchronization acquisition time in the homogeneous channel environment as compared with the another method.
The relation between the threshold and the false alarm probability is provided below. As well known to one skilled in the art, in case that a high threshold is set to reduce the false alarm probability, the synchronization detection probability lowers. In contrast, in case that a low threshold is set to increase the synchronization detection probability, the high synchronization detection probability is obtained but the false alarm probability also increases.
Therefore, it is of great importance to maintain the false alarm probability below a preset value and maximize the synchronization detection probability.
FIG. 1 is a block diagram of a conventional signal acquisition apparatus having a constant false alarm probability. Referring to FIG. 1, the conventional signal acquisition apparatus includes a matched filter 100, a received power estimator 110, an absolute value calculator 120, a multiplier 130, and a comparator 140. A received analog signal passes through an analog-to-digital converter (not shown), and is input to the matched filter 100 and the received power estimator 110.
The matched filter 100 outputs the correlation value as explained earlier, and the output value of the matched filter 100 is fed to the absolute value calculator 120 to obtain its absolute value. As a result, a magnitude of the correlation value is output.
For instance, provided that the received digital signal is ri, the output value of the matched filter 100 can be expressed as Equation 2.
                              output          ⁢                                          ⁢          of          ⁢                                          ⁢          the          ⁢                                          ⁢          matched          ⁢                                          ⁢          filter                =                              ∑                          i              =              0                                      N              -              1                                ⁢                                    r              i                        ⁢                          c                              N                -                1                            *                                                          [                  Equation          ⁢                                          ⁢          2                ]            
Let the absolute value calculated from the output of the matched filter 100 be Ck, and then Ck can be expressed as Equation 3.
                              C          k                =                                                                        ∑                                  i                  =                  0                                                  N                  -                  1                                            ⁢                                                r                  i                                ⁢                                  c                                      N                    -                    1                                    *                                                                          2                                    [                  Equation          ⁢                                          ⁢          3                ]            
The resultant value Y in intervals from 0 to M−1 with the burst size M can be expressed as Equation 4.
                    Y        =                              ∑                          k              =              0                                      M              -              1                                ⁢                      C            k                                              [                  Equation          ⁢                                          ⁢          4                ]            
The received power estimator 110 measures a power with respect to the received signal. The received power Z measured from the received signal ri can be expressed as Equation 5.
                    Z        =                              ∑                          i              =              0                                      N              -              1                                ⁢                                                                  r                i                                                    2                                              [                  Equation          ⁢                                          ⁢          5                ]            
As shown in Equation 5, the output value of the received power estimator 110 is multiplied with a constant T at the multiplier 130, and the resultant value is input to the comparator 140. The constant T is a scaling value to regulate to a target false alarm probability.
The comparator 140 compares the output value of the absolute value calculator 120 in Equation 4, with the product of the output of the received power estimator 110 in Equation 5 and the constant T. According to a result of the comparison, when the output value of the absolute value calculator 120 is greater than the output value of the multiplier 130, the synchronization acquisition is determined.
FIG. 2 and FIG. 3 are graphs showing the output value of the CFAR detector of FIG. 1.
FIG. 2 is a graph showing the output value of the conventional signal acquisition apparatus having the constant false alarm rate. Referring to FIG. 2, the magnitude 201 of the output of the matched filter 100 of FIG. 1 generates a first peak 203 by packets. Accordingly, the threshold 202 is modified based on the output of the received power estimator 110 to thus effectively detect the synchronization.
FIG. 3 is an enlarged graph of FIG. 2 with respect to one packet. In FIG. 3, as discussed early, second peaks are generated in vicinity of the first peak 203 due to the influence of the channel conditions such as noise and multipath. Also, other second peaks 301 and 302 are generated away from the first peak 203.
The second peaks around the first peak 203 mostly result from the multipath. It can be seen that the synchronization is detected at the first peak 203. Because of the second peaks 301 and 302 generated away from the first peak 203 due to the noise, the synchronization may be determined, that is, may be falsely alarmed based on the low threshold 202, though it is not actual synchronization point.
As discussed above, the increase of the threshold 202 reduces the false alarm rate but lowers the signal detection probability. Conversely, the decrease of the threshold 202 increases not only the signal detection probability but also the false alarm rate because of the second peaks 301 and 302, where the substantial synchronization is not conducted, generated away from the first peak 203.
If the threshold is determined using a function of a signal to noise ratio (SNR) of the received signal, optimal result can be attained but is infeasible in practice. Hence, the threshold is adaptively determined using the power of the received signal as in the CFAR detection, and then characteristics of the CFAR can be obtained with all SNR.
Generally, the false alarm far more affects the performance of the synchronization acquisition when the signal is present with the noise, rather than merely the noise is present. Since the detection probability is low as for the low SNR, the false alarm rate is less affected. As for the high SNR, the false alarm may be generated at a wrong location because the detection probability is almost 1. As a result, disadvantageously, the detection probability performance is degraded.
To mitigate the effects of the false alarm, verification procedure is conducted after the packet detection to double-dwell on the signal. This requires a double signal acquisition time. By doing this, the false alarm due to the noise can be reduced but the overall detection probability may fall when the detection probability in each interval is not 1.
As the packet error rate (PER) results from not only the bit error but also the false alarm in the packet detection, a new method is demanded to reduce the false alarm rate while the detection probability is maintained.